Current detection circuit and switching power supply

ABSTRACT

A current detection circuit of the present invention has a switching device  1;  an auxiliary switching device  2;  an offset voltage source comprising an offset resistor device  7  and a current source circuit  8;  a compensation circuit, comprising a differential amplifier  4  and a compensation transistor  5,  for adjusting the output current of the auxiliary switching device  2  so that the potential obtained by subtracting the voltage drop across the offset resistor device  7  from the output potential of the switching device  1  is equal to the output potential of the auxiliary switching device  2,  being configured that the detection level of the current flowing in the switching device  1  is shifted by an offset amount.

BACKGROUND OF THE INVENTION

The present invention relates to a current detection circuit fordetecting the current flowing in a switching device, such as a MIS FET(metal insulator semiconductor field effect transistor), and a switchingpower supply using the same.

In recent years, switching power supplies have been used in variouselectronic apparatuses because of their high efficiency and high powerconversion characteristics. In particular, in the case that quicktransient response characteristics are requested, the current modecontrol system, which is not affected by the resonance frequency of aninductor and an output capacitor constituting such a switching powersupply, is used. The current mode control system detects the currentflowing in the inductor and controls the current to carry out outputcontrol, and requires a current detection circuit. In the case that acurrent detection device, such as a resistor, is used in a currentdetection circuit, a conduction loss occurs and its efficiency isreduced. Hence, a current detection circuit for detecting the currentflowing in a switching device, such as a MOS FET (metal-oxidesemiconductor field effect transistor), has been proposed as describedin International Publication No. WO00/079682, for example.

FIG. 11 is a circuit configuration diagram showing a current detectioncircuit described as a conventional current detection circuit inInternational Publication No. WO00/079682. In FIG. 11, an outputtransistor 101 is formed of an N-channel MOS FET. The drain of thisoutput transistor 101 is connected to a DC power supply 110, and thesource thereof is connected to one terminal of a load 103. The otherterminal of the load 103 is grounded. An auxiliary transistor 102 is anN-channel MOS FET, the drain thereof is connected to the DC power supply110, and the source thereof is connected to the source of a compensationtransistor 105. The ratio I101/I102 of the source current I101 of theoutput transistor 101 to the source current I102 of the auxiliarytransistor 102 is designed to have a substantially constant value(hereafter, n=I101/I102) in the case that the potentials of the drain,gate and source of one of the two switching devices are made equal tothose of the other, respectively. This can be realized by making thegate length of the output transistor 101 equal to that of the auxiliarytransistor 102 and by setting the ratio of the gate width of the outputtransistor 101 to that of the auxiliary transistor 102 at n:1 in thecase that the current detection circuit is produced so as to be built ina monolithic integrated circuit, for example. The compensationtransistor 105 is a P-channel MOS FET, and the drain thereof isconnected to one terminal of a current detection resistor 106. The otherterminal of the current detection resistor 106 is grounded. Adifferential amplifier 104 detects the potential difference between theconnection point P of the output transistor 101 and the load 103 and theconnection point Q of the auxiliary transistor 102 and the compensationtransistor 105. The differential amplifier 104 amplifies the detectedpotential difference and outputs the amplified voltage to the gate ofthe compensation transistor 105. A drive circuit 111 outputs a commondrive signal to the gates of the output transistor 101 and the auxiliarytransistor 102.

The operation of the conventional current detection circuit shown inFIG. 11 will be described below.

First, when the potential at the connection point P with respect to thepotential at the connection point Q increases in the positive directionby virtue of the differential amplifier 104 and the compensationtransistor 105, the potential at the connection point Q rises.Conversely, when the potential at the connection point P increases inthe negative direction, the potential at the connection point Q lowers.Hence, the source potential (the potential at the connection point P) ofthe output transistor 101 becomes substantially equal to the sourcepotential (the potential at the connection point Q) of the auxiliarytransistor 102. Furthermore, the drain potential and the gate potentialof the output transistor 101 are equal to those of the auxiliarytransistor 102, respectively, as clearly understood from the circuitconfiguration. Hence, the potentials of the drain, gate and source ofthe output transistor 101 are equal to those of the auxiliary transistor102, respectively. The ratio I101/I102 of the source current I101 of theoutput transistor 101 and the source current I102 of the auxiliarytransistor 102 is thus maintained at the constant value n. In otherwords, the source current I102 of the auxiliary transistor 102 isI102=I101/n, a detection voltage Vs proportionate to the source currentI101 of the output transistor 101 is generated across the currentdetection resistor 106. When it is assumed that the resistance value ofthe current detection resistor 106 is Rs, the detection voltage Vs isVs=Rs·I101/n.

In the case that the conventional current detection circuit configuredas described above is applied to the detection of the current flowing ina switching device in a switching power supply, an inductor and arectifier circuit are connected to a load. In the case of a step-downconverter wherein a synchronous rectifier circuit is used in a rectifiercircuit, the current of the switching device is passed in the reversedirection in some occasions so that power is regenerated from the outputto the input in order to promptly suppress overshoots occurred in theoutput, for example. However, the conventional current detection circuitconfigured as described above can detect only the current flowing fromthe output transistor to the load. In other words, the conventionalcurrent detection circuit configured as described above cannot detectthe reverse current flowing in the switching device of the switchingpower supply.

SUMMARY OF THE INVENTION

The present invention is intended to provide a current detection circuitcapable of detecting the reverse current flowing in a switching deviceand to provide a switching power supply, the transient responsecharacteristics of which are improved by using this current detectioncircuit.

For the purpose of attaining the above-mentioned object, a currentdetection circuit in accordance with a first aspect of the presentinvention is a current detection circuit for detecting the currentflowing in a switching device. The current detection circuit includes:

-   -   an auxiliary switching device being provided in parallel with        the switching device and being ON at least when the switching        device is ON,    -   an offset voltage source for generating an offset voltage, and    -   a compensation circuit for adjusting the output current of the        auxiliary switching device so that the potential obtained by        adding the offset voltage to the output potential of the        switching device becomes equal to the output potential of the        auxiliary switching device. The current detection circuit in        accordance with the present invention configured as described        above can accurately detect the reverse current flowing in the        switching device.

A switching power supply in accordance with another aspect of thepresent invention includes a switching device, an inductor connected tothe switching device, a rectifier circuit for rectifying the output ofthe inductor, smoothing means for smoothing the current flowing in therectifier circuit, a current detection circuit for detecting the currentof the switching device, and a control circuit, to which the currentdetection signal from the current detection circuit and the outputvoltage output from the smoothing means are input, for turning ON/OFFthe switching device to control the output voltage.

Further, the current detection circuit includes:

-   -   an auxiliary switching device being provided in parallel with        the switching device and being ON at least when the switching        device is ON.    -   an offset voltage source for generating an offset voltage, and    -   a compensation circuit for adjusting the output current of the        auxiliary switching device so that the potential obtained by        adding the offset voltage to the output potential of the        switching device becomes equal to the output potential of the        auxiliary switching device. The switching power supply in        accordance with the present invention configured as described        above has excellent transient response characteristics by using        the current detection circuit capable of detecting the reverse        current flowing in the switching device.

The current detection circuit in accordance with the present inventioncan surely detect the reverse current flowing in the switching device byusing a simple configuration.

In addition, the reverse current of the switching device, flowing fromthe output to the input so that power is regenerated, can be detectedand controlled by applying the current detection circuit in accordancewith the present invention to the detection of the current flowing inthe switching device of the switching power supply conforming to thecurrent mode control system. Hence, overshoots occurred in the outputvoltage owing to abrupt reduction in the output current, for example,can be suppressed.

Furthermore, in the switching power supply using the current detectioncircuit in accordance with the present invention and conforming to thecurrent mode control system, the change width of the setting value ofthe operation current is limited to the necessary minimum during theordinary operation, and the limitation in the change width of thesetting value of the operation current is relaxed during a transientperiod, for example, when the output conditions change abruptly. Hence,the switching power supply produces excellent effects of quickeningresponsiveness and smoothing return to the ordinary operation.

Still further, in the switching power supply using the current detectioncircuit in accordance with the present invention and conforming to thecurrent mode control system, the offset level of the current detectioncircuit is made variable. During a transient period, for example, whenthe output conditions change abruptly, the offset level is set at alarge value, whereby the detection level is shifted. Hence, theswitching power supply has excellent effects of quickeningresponsiveness and smoothing return to the ordinary operation.

While the novel features of the invention are set forth particularly inthe appended claims, the invention, both as to configuration andcontent, will be better understood and appreciated, along with otherobjects and features thereof, from the following detailed descriptiontaken in conjunction with the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram showing the configuration of a currentdetection circuit in accordance with a first embodiment of the presentinvention;

FIG. 2 is a characteristic graph in the current detection circuit inaccordance with the first embodiment of the present invention;

FIG. 3 is a circuit diagram showing the configuration of a currentdetection circuit in accordance with a second embodiment of the presentinvention;

FIG. 4 is a circuit diagram showing the configuration of a currentdetection circuit in accordance with a third embodiment of the presentinvention;

FIG. 5 is a circuit diagram showing the configuration of a switchingpower supply in accordance with a fourth embodiment of the presentinvention;

FIG. 6 is a waveform diagram showing operations in the switching powersupply in accordance with the fourth embodiment of the presentinvention;

FIG. 7 is a circuit diagram showing the configuration of a switchingpower supply in accordance with a fifth embodiment of the presentinvention;

FIG. 8 is a characteristic graph in the current detection circuit of theswitching power supply in accordance with the fifth embodiment of thepresent invention;

FIG. 9 is a circuit diagram showing the configuration of a switchingpower supply in accordance with a sixth embodiment of the presentinvention;

FIG. 10 is a characteristic graph in the current detection circuit ofthe switching power supply in accordance with the sixth embodiment ofthe present invention; and

FIG. 11 is the circuit diagram showing the configuration of theconventional current detection circuit.

It will be recognized that some or all of the Figures are schematicrepresentations for purposes of illustration and do not necessarilydepict the actual relative sizes or locations of the elements shown.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Preferred embodiments of a current detection circuit and a switchingpower supply using the same in accordance with the present inventionwill be described below referring to the accompanying drawings.

First Embodiment

FIG. 1 is a circuit diagram showing the configuration of a currentdetection circuit in accordance with a first embodiment of the presentinvention. In the current detection circuit in accordance with the firstembodiment, an output transistor 1 is a switching device and formed ofan N-channel MIS FET (metal insulator semiconductor field effecttransistor). An auxiliary transistor 2, connected in parallel with theoutput transistor 1, is formed of an N-channel MIS FET. In addition, acompensation transistor 5 is formed of a P-channel MIS FET. In FIG. 1,numeral 3 designates a load, numeral 4 designates a differentialamplifier, numeral 6 designates a current detection resistor, numeral 7designates an offset resistor device, numeral 8 designates a currentsource circuit, numeral 9 designates a drive circuit, and numeral 10designates a DC power supply.

The drain of the output transistor 1 is connected to the DC power supply10, the gate thereof is connected to the drive circuit 9, and the sourcethereof is connected to the load 3. The drain of the auxiliarytransistor 2 is connected to the DC power supply 10, the gate thereof isconnected to the drive circuit 9, the source thereof is connected to thesource of the compensation transistor 5. One terminal of the load 3 isconnected to the source of the output transistor 1, and the otherterminal of the load 3 is grounded. A current is supplied from thesource of the output transistor 1 being ON to the load 3 connected asdescribed above. The ratio I1/I2 of the source current I1 of the outputtransistor 1 and the source current I2 of the auxiliary transistor 2 isdesigned to have a substantially constant value (hereafter, n=I1/I2) inthe case that the potentials of the drain, gate and source of one of thetwo transistors are made equal to those of the other, respectively. Theratio of the size of the output transistor 1 to that of the auxiliarytransistor 2 is set at n:1 in the case that the current detectioncircuit is produced so as to be built in a monolithic integratedcircuit, for example. Setting the size ratio at n:1 is to make the gatelengths of the two transistors equal and to set the ratio of the gatewidths of the two transistors at n:1, for example. The drain of thecompensation transistor 5 is connected to one terminal of the currentdetection resistor 6. The other terminal of the current detectionresistor 6 is grounded.

One terminal of the offset resistor device 7 is connected to theconnection point of the output transistor 1 and the load 3. To the otherterminal of the offset resistor device 7, one terminal of the currentsource circuit 8 is connected. The other terminal of the current sourcecircuit 8 is grounded. One terminal (the negative input terminal) of theinput terminals of the differential amplifier 4 is connected to theconnection point Q of the auxiliary transistor 2 and the compensationtransistor 5. The other input terminal (the positive input terminal) ofthe differential amplifier 4 is connected to the connection point P ofthe offset resistor device 7 and the current source circuit 8. Thedifferential amplifier 4 detects the potential difference between theconnection point P and the connection point Q. The differentialamplifier 4 amplifies the detected potential difference and outputs theamplified voltage to the gate of the compensation transistor 5.

The drive circuit 9 outputs a common drive signal to the outputtransistor 1 and the auxiliary transistor 2, thereby turning ON/OFF theoutput transistor 1 and the auxiliary transistor 2. In the firstembodiment, the H level drive signal output from the drive circuit 9 hasa high potential capable of sufficiently turning ON the outputtransistor 1 and the auxiliary transistor 2. Furthermore, the L leveldrive signal output from the drive circuit 9 has a low potential capableof sufficiently turning OFF the output transistor 1 and the auxiliarytransistor 2.

The operation of the current detection circuit in accordance with thefirst embodiment shown in FIG. 1 will be described below.

In the current detection circuit in accordance with the first embodimentshown in FIG. 1, when the potential at the connection point P withrespect to the potential at the connection point Q increases in thepositive direction by virtue of the differential amplifier 4 and thecompensation transistor 5, the potential at the connection point Qrises. Conversely, when the potential at the connection point P withrespect to the potential at the connection point Q increases in thenegative direction, the potential at the connection point Q lowers.Hence, the potential at the connection point P of the offset resistordevice 7 and the current source circuit 8 becomes substantially equal tothe source potential (the potential at the connection point Q) of theauxiliary transistor 2.

On the other hand, the ratio of the ON resistance of the outputtransistor 1 to the ON resistance of the auxiliary transistor 2, the twotransistors being turned ON by the drive circuit 9, is substantiallyinversely proportionate to the above-mentioned size ratio (n:1). Inother words, when it is assumed that the ON resistance of the outputtransistor 1 is Ron, the ON resistance of the auxiliary transistor 2 isn·Ron. Hence, when it is assumed that the potential of the DC powersupply 10 is Vi and that the source current of the output transistor 1is I1, the source potential of the output transistor 1 is (Vi−I1·Ron).Since the potential at the connection point P is obtained by subtractingthe voltage drop across the offset resistor device 7 from the sourcepotential of the output transistor 1, when it is assumed that theresistance value of the offset resistor device 7 is Rx and that thecurrent value of the current source circuit 8 is Ix, the potential atthe connection point P is (Vi−I1·Ron−Ix·Rx). When it is assumed that thesource current of the auxiliary transistor 2 is I2, the potential at theconnection point Q is (Vi−n·Ron·I2). Since the potential at theconnection point Q becomes equal to the potential at the connectionpoint P, the source current I2 of the auxiliary transistor 2 isrepresented by the following expression (1).I2=(I1+Ix·Rx/Ron)/n (1)

When it is assumed that the resistance value of the current detectionresistor 6 is Rs, a detection voltage Vs being related to the sourcecurrent I1 of the output transistor 1 as represented by the followingexpression (2) is generated across the current detection resistor 6.Vs=Rs·(I1+Ix·Rx/Ron)/n   (2)

FIG. 2 is a characteristic graph of the current detection circuit inaccordance with the first embodiment, showing the relationship of theabove-mentioned expression (2). When the current value Ix of the currentsource circuit 8 is zero (Ix=0), the expression (2) is Vs=Rs·I1/n. Thedetection voltage Vs is proportionate to the source current I1 of theoutput transistor 1. In this case, the source current I1 can be detectedonly in the positive direction.

In the case that the current value Ix of the current source circuit 8 islarger than zero (Ix>0), the detection voltage Vs is (Rs·Ix·Rx/Ron)/neven when the source current I1 of the output transistor 1 is zero.Furthermore, even when the source current I1 is negative, detection ispossible in the range of the source current I1 reaching the value of(−Ix·Rx/Ron) at which Vs=0.

In the case that the current value Ix of the current source circuit 8 issmaller than zero (Ix<0), that is, in the case that the current of thecurrent source circuit 8 flows in the offset resistor device 7 in thereverse direction, the value of (−Ix·Rx/Ron) at which Vs=0 is positive.In this case, the source current I1 of the output transistor 1 havinglarger output can be detected.

As described above, in the current detection circuit in accordance withthe first embodiment of the present invention, the offset is added tothe detection current by providing the offset resistor device 7 and thecurrent source circuit 8. With this configuration, it is possible todetect the source current I1 of the output transistor 1 even when thecurrent flows in the negative direction.

In the current detection circuit in accordance with the firstembodiment, the offset resistor device 7 and the current source circuit8 constitute an offset voltage source. Furthermore, the differentialamplifier 4 and the compensation transistor 5 constitute a compensationcircuit.

Second Embodiment

A current detection circuit in accordance with a second embodiment ofthe present invention will be described below using the accompanyingdrawing, FIG. 3. FIG. 3 is a circuit diagram showing the configurationof the current detection circuit in accordance with the secondembodiment.

In FIG. 3, numeral 1A designates an output transistor serving as aswitching device and formed of a P-channel MIS FET. Numeral 2Adesignates an auxiliary transistor formed of a P-channel MIS FET.Numeral 3 designates a load, numeral 4 designates a differentialamplifier, numeral 5 designates a compensation transistor formed of aP-channel MIS FET, numeral 6 designates a current detection resistor,numeral 7A designates an offset resistor device formed of a P-channelMIS FET, numeral 8 designates a current source circuit, numeral 9designates a drive circuit, and numeral 10 designates a DC power supply.The configuration of the current detection circuit in accordance withthe second embodiment differs from that of the current detection circuitin accordance with the first embodiment shown in FIG. 1 described abovein that the output transistor 1A, the auxiliary transistor 2A and theoffset resistor device 7A are each formed of a P-channel MIS FET andthat the gate of the auxiliary transistor 2A and the gate of the offsetresistor device 7A are grounded. Since the other components are similarto those of the first embodiment shown in FIG. 1, their detailedconfigurational descriptions are omitted.

In FIG. 3, the source of the output transistor 1A is connected to the DCpower supply 10, the gate thereof is connected to the drive circuit 9A,and the drain thereof is connected to the load 3. The source of theauxiliary transistor 2A is connected to the DC power supply 10, the gatethereof is grounded, and the drain thereof is connected to the source ofthe compensation transistor 5. The source of the offset resistor device7A is connected to the connection point of the output transistor 1A andthe load 3, the drain thereof is connected to the current source circuit8, and the gate thereof is grounded.

The current detection circuit in accordance with the second embodimentis produced so as to be built in a monolithic integrated circuit, andthe ratio of the size of the output transistor 1A, the size of theauxiliary transistor 2A and the size of the offset resistor device 7A isset at n:1:k. Setting the size ratio at n:1:k is to make the gatelengths of the three transistors equal and to set the ratio of the gatewidths of the three transistors at n:1:k, for example.

The differential amplifier 4 detects the potential difference betweenthe connection point P of the offset resistor device 7A and the currentsource circuit 8 and the connection point Q of the auxiliary transistor2A and the compensation transistor 5, and amplifies the potentialdifference. The differential amplifier 4 then outputs the amplifiedvoltage to the gate of the compensation transistor 5. The drive circuit9A outputs a drive signal to the gate of the output transistor 1,thereby turning ON/OFF the output transistor 1A. The L level drivesignal output from the drive circuit 9A has a ground potential capableof sufficiently turning ON the output transistor 1A. Furthermore, the Hlevel drive signal output from the drive circuit 9A has a high potential(for example, the power supply voltage Vi) capable of sufficientlyturning OFF the output transistor 1A.

The operation of the current detection circuit in accordance with thesecond embodiment shown in FIG. 3 will be described below.

In the current detection circuit in accordance with the secondembodiment, when the potential at the connection point P with respect tothe potential at the connection point Q increases in the positivedirection by virtue of the differential amplifier 4 and the compensationtransistor 5, the potential at the connection point Q rises as in thecase of the above-mentioned first embodiment. Conversely, when thepotential at the connection point P with respect to the potential at theconnection point Q increases in the negative direction, the potential atthe connection point Q lowers. Hence, the potential at the connectionpoint P of the offset resistor device 7A and the current source circuit8 becomes substantially equal to the drain potential (the potential atthe connection point Q) of the auxiliary transistor 2A.

The auxiliary transistor 2A is ON at all times since the gate thereof isgrounded. The ratio of the ON resistance of the output transistor 1Abeing turned ON by the drive circuit 9A to the ON resistance of theauxiliary transistor 2A is substantially inversely proportionate to theratio (n:1:k) of the sizes of the output transistor 1A, the auxiliarytransistor 2A and the offset resistor device 7A described above. Inother words, when it is assumed that the ON resistance of the outputtransistor 1A is Ron, the ON resistance of the auxiliary transistor 2Ais n·Ron. Furthermore, the offset resistor device 7A is also ON sincethe gate thereof is grounded, whereby the resistance value thereof is(n/k)·Ron. Hence, when it is assumed that the potential of the DC powersupply 10 is Vi and that the drain current of the output transistor 1Ais I1A, the drain potential of the output transistor 1A is (Vi−I1A·Ron).Since the potential at the connection point P is obtained by subtractingthe voltage drop across the offset resistor device 7A from the drainpotential of the output transistor 1A, when it is assumed that thecurrent value of the current source circuit 8 is Ix, the potential atthe connection point P is (Vi−I1A·Ron−Ix·(n/k)·Ron). When it is assumedthat the drain current of the auxiliary transistor 2A is I2A, thepotential at the connection point Q is (Vi−n·Ron·I2A). Since thepotential at the connection point Q becomes equal to the potential atthe connection point P, the drain current I2A of the auxiliarytransistor 2A is represented by the following expression (3).I2A=I1A/n+Ix/k   (3)

When it is assumed that the resistance value of the current detectionresistor 6 is Rs, a detection voltage Vs being related to the draincurrent I1A of the output transistor 1A as represented by the followingexpression (4) is generated in the current detection resistor 6.Vs=Rs·(I1A/n+Ix/k)   (4)

As described above, in the current detection circuit accordance with thesecond embodiment of the present invention, the detection current isoffset by providing the offset resistor device 7A and the current sourcecircuit 8. With this configuration, it is possible to detect the draincurrent I1A of the output transistor 1A even when the current flows inthe negative direction. Furthermore, like the output transistor 1A, theoffset resistor device 7A is formed of a P-channel MIS FET so that theON resistance thereof has a value corresponding to the size ratio in theON state. Hence, the influence of the temperature characteristics andmanufacturing fluctuations of the ON resistance can be eliminated. Forthis reason, the current detection circuit in accordance with the secondembodiment of the present invention can carry out highly accuratecurrent detection.

Third Embodiment

A current detection circuit in accordance with a third embodiment of thepresent invention will be described below using the accompanyingdrawing, FIG. 4. FIG. 4 is a circuit diagram showing the configurationof the current detection circuit in accordance with the third embodimentof the present invention.

As shown in FIG. 4, in the current detection circuit in accordance withthe third embodiment, a load 3B is connected to the power supplypotential side. An output transistor 1B serving as a switching device isformed of an N-channel MIS FET, the source of which is grounded. In FIG.4, an auxiliary transistor 2B, a compensation transistor 5B and anoffset resistor device 7B are all formed of an N-channel MIS FET. Thegates of the auxiliary transistor 2B and the offset resistor device 7Bhave the power supply potential Vi, and the source of the auxiliarytransistor 2B is grounded. A current source circuit 8B is configured topass a current to the offset resistor device 7B, and a current detectionresistor 6B is connected to the power supply potential side.

As described above, the current detection circuit in accordance with thethird embodiment has a configuration wherein the connections to thepower supply potential side and the ground potential side in the currentdetection circuit in accordance with the second embodiment shown in FIG.3 described above are reversed. Hence, the current detection circuit inaccordance with the third embodiment is equivalent to the currentdetection circuit in accordance with the second embodiment except thatthe current direction and the voltage polarity are inverted. For thisreason, the current detection circuit in accordance with the thirdembodiment of the present invention configured as described above canalso carry out highly accurate current detection.

Fourth Embodiment

A switching power supply in accordance with a fourth embodiment of thepresent invention will be described below using the accompanyingdrawings, FIGS. 5 and 6. FIG. 5 is a circuit diagram showing theconfiguration of the switching power supply in accordance with thefourth embodiment. FIG. 6 is a waveform diagram showing the operationsof the main sections of the switching power supply in accordance withthe fourth embodiment. The switching power supply in accordance with thefourth embodiment is a switching power supply having the configurationof the current detection circuit in accordance with the secondembodiment described above

In FIG. 5, the switching power supply in accordance with the fourthembodiment is provided with a DC power supply 10 for outputting a powersupply voltage Vi and a step-down converter 20, and supplies an outputcurrent Io to a load 32. The step-down converter 20 comprises aswitching device 21 formed of a P-channel MIS FET, an auxiliarytransistor 22 formed of a P-channel MIS FET, a synchronous rectifiertransistor 23 formed of an N-channel MIS FET, a differential amplifier24, a compensation transistor 25 formed of a P-channel MIS FET, acurrent detection resistor 26 having a resistance value Rs, an offsetresistor device 27 formed of a P-channel MIS FET, a current sourcecircuit 28, a control circuit 29 for driving the switching device 21 andthe synchronous rectifier transistor 23, an inductor 30, and an outputcapacitor 31. The synchronous rectifier transistor 23 serves as arectifier circuit, and the output capacitor 31 serves as a smoothingmeans.

In the switching power supply in accordance with the fourth embodiment,at least the switching device 21, the auxiliary transistor 22 and theoffset resistor device 27 are produced so as to be built in a monolithicintegrated circuit, and the ratio of the size of the switching device21, the size of the auxiliary transistor 22 and the size of the offsetresistor device 27 is set at n:1:k. The switching device 21 in theswitching power supply in accordance with the fourth embodimentcorresponds to the output transistor 1A in the current detection circuitin accordance with the second embodiment described above.

The detection voltage Vs from the current detection resistor 26 is inputto the control circuit 29, and the control circuit 29 detects the outputvoltage Vo supplied from the output capacitor 31 to the load 32. Then,the control circuit 29 sets the upper limit value of the detectionvoltage Vs so as to stabilize the output voltage Vo and turns ON/OFF theswitching device 21 and the synchronous rectifier transistor 23alternately.

In the waveform diagram of FIG. 6, (a) shows an output current Iosupplied from the output capacitor 31 to the load 32, (b) shows acurrent IL flowing in the inductor 30, (c) shows the detection voltageVs and the upper limit setting value Vc of the detection voltage Vs, thevalue being set by the control circuit 29, and (d) shows the outputvoltage Vo.

The operation of the switching power supply in accordance with thefourth embodiment will be described below referring to FIG. 6, thewaveform diagram showing the operations of the main sections.

First, the ordinary operation in the left portion of the waveformdiagram of FIG. 6 will be described below.

When the synchronous rectifier transistor 23 is turned OFF and theswitching device 21 is turned ON by a drive signal from the controlcircuit 29 at time t0, a current flows in the inductor 30 from the DCpower supply 10 via the switching device 21. This inductance current ILincreases in proportion to the input/output voltage difference (Vi−Vo)of the step-down converter 20 with time, the input/output voltagedifference being applied to the inductor 30. The auxiliary transistor22, the differential amplifier 24, the compensation transistor 25, thecurrent detection resistor 26, the offset resistor device 27 and thecurrent source circuit 28 are similar in configuration to those of thecurrent detection circuit in accordance with the second embodiment. Thedetection voltage Vs represented by the following expression (5) isgenerated across the current detection resistor 26.Vs=Rs (IL/n+Ix/k)   (5)

The control circuit 29 detects the output voltage Vo, sets the upperlimit setting value Vc of the detection voltage Vs so as to stabilizethe output voltage Vo, and compares the detection voltage Vs with theupper limit setting value Vc. When the detection voltage Vs reaches theupper limit setting value Vc at time t1, the control circuit 29 turnsOFF the switching device 21 and then turns ON the synchronous rectifiertransistor 23. The inductance current IL decreases in proportion to theoutput voltage Vo serving as the voltage applied to the inductor 30. Thecontrol circuit 29, having an oscillator (not shown) therein, turns OFFthe synchronous rectifier transistor 23 and turns ON the switchingdevice 21 at time t2. The control circuit 29 repeats the above-mentionedoperation. In the case that the upper limit setting value Vc of thedetection voltage Vs is set at a higher value, the ON state of theswitching device 21 becomes longer, the peak value of the inductancecurrent IL becomes larger, and the current for charging the outputcapacitor 31 increases. Conversely, in the case that the upper limitsetting value Vc is set at a lower value, the peak value of theinductance current IL becomes smaller, and the current for charging theoutput capacitor 31 decreases. In other words, the output power can becontrolled by changing the setting of the upper limit setting value Vc.A control method for stabilizing the output voltage Vo by controllingthe inductance current IL as described above is referring to as thecurrent mode control system.

Next, the operation in the case that the output current Io has decreasedabruptly to zero at time t3 in FIG. 6 will be described below.

After the output current Io decreases abruptly, the power supply fromthe step-down converter 20 to the load 32 becomes excessive, and theoutput voltage Vo rises. The control circuit 29 lowers the upper limitsetting value Vc of the detection voltage Vs to reset the raised outputvoltage Vo. As the upper limit setting value Vc lowers, the ON time ofthe switching device 21 becomes shorter, and the inductance current ILdecreases. At time t4, the inductance current IL is zero, and thedetection voltage Vs is Rs·Ix/k. The output voltage Vo stops rising atthis time. After time t4, the inductance current IL is lower than zero.Then, at time t5, the peak value of the inductance current IL decreasesuntil −(n/k)·Ix corresponding to the value obtained at Vc=0, whereby theoutput voltage Vo is kept on lowering. When the output voltage Vo comesclose to a target value at time t6, the control circuit 29 raises theupper limit setting value Vc, thereby braking the lowering of the outputvoltage Vo. Then, the output voltage Vo reaches the target value whilebeing subjected to damped oscillation.

As described above, the switching power supply in accordance with thefourth embodiment of the present invention conforms to the current modecontrol system capable of controlling the current flowing even in thenegative direction. Hence, in the case that the output voltage risesowing to abrupt decrease in load, for example, the inductance current ispassed reversely to quickly lower the raised output voltage. Therefore,the switching power supply in accordance with the fourth embodiment canproduce excellent effects of suppressing the generation of overshoots ofthe output voltage and promptly returning the output voltage to thetarget value.

In the switching power supply in accordance with the fourth embodiment,the offset resistor device 27 and the current source circuit 28constitute an offset voltage source. Furthermore, the differentialamplifier 24 and the compensation transistor 25 constitute acompensation circuit.

Fifth Embodiment

A switching power supply in accordance with a fifth embodiment of thepresent invention will be described below using the accompanyingdrawings, FIGS. 7 and 8. FIG. 7 is a circuit diagram showing theconfiguration of the switching power supply in accordance with the fifthembodiment. FIG. 8 is a characteristic graph showing the relationshipbetween the inductance current IL and the detection voltage Vs in theswitching power supply in accordance with the fifth embodiment. In FIG.7, the components having the same functions and configurations as thoseof the components of the switching power supply in accordance with thefourth embodiment shown in FIG. 5 are designated by the same numerals,and their descriptions are omitted. The switching power supply inaccordance with the fifth embodiment differs from the switching powersupply in accordance with the fourth embodiment in the configuration ofthe control circuit. In other words, a function capable of coping withabrupt change in the output voltage is added to the control circuit 29in accordance with the fourth embodiment. Hence, the control circuit inaccordance with the fifth embodiment is referred to as “control circuit29A” so as to be distinguished from the control circuit 29 in accordancewith the fourth embodiment shown in FIG. 5. The internal configurationof the control circuit 29A will be described below specifically.

In FIG. 7, the control circuit 29A comprises a reference voltage source290, an output voltage detection circuit 291, an error amplifier 292, afirst comparator 293, a second comparator 294, a first clamp circuit295, a second clamp circuit 296, a clock circuit 297, a currentcomparator 298, an RS latch 299, and a drive circuit 300. The referencevoltage source 290 is variable. The reference voltage Vr of thereference voltage source 290 is applied to the positive input terminalof the error amplifier 292, the negative input terminal of the firstcomparator 293 and the positive input terminal of the second comparator294. The output voltage detection circuit 291 comprises resistors 2911to 2914 connected in series and divides the output voltage Vo. Theconnection point of the resistor 2911 and the resistor 2912 is connectedto the positive input terminal of the first comparator 293. Theconnection point of the resistor 2912 and the resistor 2913 is connectedto the negative input terminal of the error amplifier 292. Theconnection point of the resistor 2913 and the resistor 2914 is connectedto the negative input terminal of the second comparator 294.

The error amplifier 292 outputs the upper limit setting value Vc, andthe output of the error amplifier 292 is connected to the negative inputterminal of the current comparator 298. The output of the firstcomparator 293 is input to the first clamp circuit 295. When the outputof the first comparator 293 is H level, the first clamp circuit 295clamps the upper limit setting value Vc at a first setting voltage Vc1or less. When the output of the first comparator 293 is L level, theclamping of the upper limit setting value Vc at the first settingvoltage Vc1 or less is released. The output of the second comparator 294is input to the second clamp circuit 296. When the output of the secondcomparator 294 is H level, the second clamp circuit 296 clamps the upperlimit setting value Vc at a second setting voltage Vc2 or more. When theoutput of the second comparator 294 is L level, the clamping of theupper limit setting value Vc at the second setting voltage Vc2 or moreis released.

The clock circuit 297 outputs a clock signal having a predeterminedfrequency to the set terminal of the RS latch 299. The detection voltageVs serving as a current detection signal is input to the positive inputterminal of the current comparator 298, and the upper limit settingvalue Vc is input to the negative input terminal thereof. The currentcomparator 298 outputs the signal indicating the result of thecomparison to the reset terminal of the RS latch 299. The drive circuit300 turns ON/OFF the switching device 21 and the synchronous rectifiertransistor 23 alternately on the basis of the output of the RS latch299.

Next, the operation of the control circuit 29A of the switching powersupply in accordance with the fifth embodiment will be described below.

First, the ordinary operation will be described below. The outputvoltage Vo is divided by the output voltage detection circuit 291 andcompared with the reference voltage Vr by the error amplifier 292. Atthis time, the reference voltage Vr is set at a first voltage Vr1. Theupper limit setting value Vc of the detection voltage output from theerror amplifier 292 lowers when the output voltage Vo is higher than atarget value, and the upper limit setting value Vc rises when the outputvoltage Vo is lower than the target value. The RS latch 299 having beenset by the clock signal from the clock circuit 297 outputs an H levelsignal, whereby the drive circuit 300 turns OFF the synchronousrectifier transistor 23 and turns ON the switching device 21. As aresult, a current flows from the DC power supply 10 to the inductor 30via the switching device 21. As this inductance current IL increases,the detection voltage Vs rises.

When the detection voltage Vs becomes higher than the upper limitsetting value Vc, the output of the current comparator 298 becomes Hlevel, and the RS latch 299 having been reset outputs an L level signal.On the basis of the L level signal from the RS latch 299, the drivecircuit 300 turns OFF the switching device 21 and turns ON thesynchronous rectifier transistor 23. This state is maintained until theclock signal from the clock circuit 297 is generated. When the clocksignal from the clock circuit 297 is input to the RS latch 299, the RSlatch 299 outputs an H level signal, and the above-mentioned operationis repeated. As described above, the switching device 21 and thesynchronous rectifier transistor 23 are alternately turned ON/OFF by thedrive circuit 300. Hence, the step-down converter 20 supplies thestabilized output voltage Vo to the load 32. During this ordinaryoperation, the output voltage Vo is stabilized to the target value.Hence, the outputs of the first comparator 293 and the second comparator294 have H level, and the upper limit setting value Vc is clamped at thesecond setting voltage Vc2 or more and the first setting voltage Vc1 orless.

Next, the operation in the case that, for the purpose of abruptlyraising the output voltage Vo to another target value, the referencevoltage Vr of the reference voltage source 290 is set so as to be raisedabruptly from the first voltage Vr1 to a second voltage Vr2 will bedescribed below.

Since the reference voltage Vr has been changed from the first voltageVr1 to the second voltage Vr2, the error amplifier 292 raises the upperlimit setting value Vc to raise the output voltage Vo. At the same time,the output of the first comparator 293 has L level since the voltageapplied to the negative input terminal thereof has been raised to thesecond voltage Vr2. As a result, the clamping of the upper limit settingvalue Vc at the first setting voltage Vc1 or less is released. Hence,the upper limit setting value Vc becomes higher than the first settingvoltage Vc1, and the inductance current IL becomes larger than themaximum value obtained during the ordinary operation. Consequently, thecharging of the output capacitor 31 proceeds quickly. As the outputvoltage Vo becomes close to the target value set by the second voltageVr2 of the reference voltage source 290, the output of the firstcomparator 293 becomes H level, and the clamping of the upper limitsetting value Vc at the first setting voltage Vc1 or less is carriedout. For this reason, the upper limit setting value Vc lowers to thefirst setting voltage Vc1, and the inductance current IL is limited tothe maximum value obtained during the ordinary operation. As a result,the charging of the output capacitor 31 is braked. Eventually, the upperlimit setting value Vc becomes lower than the first setting voltage Vc1,and the output voltage Vo is stabilized to the target value set by thesecond voltage Vr2 of the reference voltage source 290.

Next, the operation in the case that, for the purpose of abruptlylowering the output voltage Vo to still another target value, thereference voltage Vr of the reference voltage source 290 is set so as tobe lowered abruptly from the second voltage Vr2 to the first voltage Vr1will be described below.

Since the reference voltage Vr has been changed from the second voltageVr2 to the first voltage Vr1, the error amplifier 292 lowers the upperlimit setting value Vc to lower the output voltage Vo. At the same time,the output of the second comparator 294 has L level since the voltageapplied to the positive input terminal thereof has been lowered to thefirst voltage Vr1. As a result, the clamping of the upper limit settingvalue Vc at the second setting voltage Vc2 or more is released. Hence,the upper limit setting value Vc becomes lower than the second settingvoltage Vc2, and the inductance current IL becomes smaller than theminimum value obtained during the ordinary operation and flowsreversely. As a result, the discharging of the output capacitor 31proceeds quickly. As the output voltage Vo becomes close to the targetvalue set by the first voltage Vr1, the output of the second comparator294 becomes H level, and the clamping of the upper limit setting valueVc at the second setting voltage Vc2 or more is carried out. For thisreason, the upper limit setting value Vc rises to the second settingvoltage Vc2, and the inductance current IL is limited to the minimumvalue obtained during the ordinary operation. As a result, thedischarging of the output capacitor 31 is braked. Eventually, the upperlimit setting value Vc becomes higher than the second setting voltageVc2, and the output voltage Vo is stabilized to the target value set bythe first voltage Vr1.

FIG. 8 is a characteristic graph showing the relationship between theinductance current IL and the detection voltage Vs of the switchingpower supply in accordance with the fifth embodiment.

As shown in FIG. 8, in the switching power supply in accordance with thefifth embodiment, the change width of the inductance current IL duringthe ordinary operation is limited in the range set using the firstsetting voltage Vc1 and the second setting value Vc2, that is, thenecessary minimum. Furthermore, when necessary, the limitation range ofthe inductance current IL is eliminated, and power supply andregeneration are carried out quickly. Therefore, the switching powersupply in accordance with the fifth embodiment is capable of quicklycoping with abrupt change in the output voltage and also capable ofcarrying out smooth return to the ordinary operation.

Sixth Embodiment

A switching power supply in accordance with a sixth embodiment of thepresent invention will be described below using the accompanyingdrawings, FIGS. 9 and 10. FIG. 9 is a circuit diagram showing theconfiguration of the switching power supply in accordance with the sixthembodiment. FIG. 10 is a characteristic graph showing the relationshipbetween the inductance current IL and the detection voltage Vs in theswitching power supply in accordance with the sixth embodiment.

In FIG. 9, the components having the same functions and configurationsas those of the components of the switching power supply in accordancewith the fourth embodiment shown in FIG. 5 are designated by the samenumerals, and their descriptions are omitted. The switching power supplyin accordance with the sixth embodiment differs from the switching powersupply in accordance with the fourth embodiment in the configuration ofthe current source circuit and the control circuit. Hence, the currentsource circuit and the control circuit in accordance with the sixthembodiment are referred to as “current source circuit 28B” and “controlcircuit 29B” so as to be distinguished from the current source circuit28 and the control circuit 29 in accordance with the fourth embodimentshown in FIG. 5. The internal configurations of the current sourcecircuit 28B and the control circuit 29B will be described belowspecifically.

The configurations of the current source circuit 28B and the controlcircuit 29B in the switching power supply in accordance with the sixthembodiment shown in FIG. 9 will be described below.

In FIG. 9, the current source circuit 28B comprises a first currentsource circuit 281 for passing a current from the drain of the offsetresistor device 27 to the ground and a second current source circuit 282for passing a current to the drain of the offset resistor device 27. Thefirst current source circuit 281 and the second current source circuit282 each have a control terminal. When an H level signal is input tothese control terminals, the first current source circuit 281 and thesecond current source circuit 282 are activated.

In the control circuit 29B in the switching power supply shown in FIG.9, the components having the same functions and configurations as thoseof the components of the control circuit 29A in accordance with thefifth embodiment shown in FIG. 7 are designated by the same numerals,and their descriptions are omitted. The control circuit 29B shown inFIG. 9 differs from the control circuit 29A shown in FIG. 7 in that afirst comparator 293B and a second comparator 294B are used instead ofthe first comparator 293 and the second comparator 294 and that thefirst clamp circuit 295 and the second clamp circuit 296 are notprovided.

In the control circuit 29B, the reference voltage Vr of the referencevoltage source 290 is applied to the positive input terminal of thefirst comparator 293B and the negative input terminal of the secondcomparator 294B. In addition, the connection point of the resistor 2911and the resistor 2912 is connected to the negative input terminal of thefirst comparator 293B. The connection point of the resistor 2913 and theresistor 2914 is connected to the positive input terminal of the secondcomparator 294B. The output V3 of the first comparator 293B is appliedto the control terminal of the first current source circuit 281, and theoutput V4 of the second comparator 294B is applied to the controlterminal of the second current source circuit 282

Next, the operation of the control circuit 29B of the switching powersupply in accordance with the sixth embodiment will be described below.

First, the ordinary operation will be described below. The outputvoltage Vo is divided by the output voltage detection circuit 291 andcompared with the reference voltage Vr of the reference voltage source290 by the error amplifier 292. The upper limit setting value Vc of thedetection voltage output from the error amplifier 292 lowers when theoutput voltage Vo is higher than a target value, and the upper limitsetting value Vc rises when the output voltage Vo is lower than thetarget value. The RS latch 299 having been set by the clock signal fromthe clock circuit 297 outputs an H level signal, whereby the drivecircuit 300 turns OFF the synchronous rectifier transistor 23 and turnsON the switching device 21. In this state, a current flows from the DCpower supply 10 to the inductor 30 via the switching device 21. As thisinductance current IL increases, the detection voltage Vs rises. Whenthe detection voltage Vs becomes higher than the upper limit settingvalue Vc, the output of the current comparator 298 becomes H level, andthe RS latch 299 having been reset outputs an L level signal. On thebasis of this L level signal, the drive circuit 300 turns OFF theswitching device 21 and turns ON the synchronous rectifier transistor23. This state is maintained until the clock signal from the clockcircuit 297 is generated.

When the clock signal from the clock circuit 297 is input to the RSlatch 299, the RS latch 299 outputs an H level signal, and theabove-mentioned operation is repeated. As described above, the switchingdevice 21 and the synchronous rectifier transistor 23 are alternatelyturned ON/OFF by the drive circuit 300. Hence, the step-down converter20 supplies the stabilized output voltage Vo to the load 32. During thisordinary operation, the output voltage Vo is stabilized to the targetvalue. Hence, the outputs of the first comparator 293B and the secondcomparator 294B have L level, whereby both the first current sourcecircuit 281 and the second current source circuit 282 of the currentsource circuit 28B are inactive. Hence, no current flows in the offsetresistor device 27, and the voltage across the terminals thereof iszero. In other words, no offset voltage is generated, and the detectionvoltage Vs generated across the current detection resistor 26 isproportionate to the current flowing in the switching device 21.

Next, the operation in the case that, for the purpose of abruptlyraising the output voltage Vo to another target value, the referencevoltage Vr is set so as to be raised abruptly from the first voltage Vr1to the second voltage Vr2 will be described below.

Since the reference voltage Vr has been changed from the first voltageVr1 to the second voltage Vr2, the error amplifier 292 raises the upperlimit setting value Vc to raise the output voltage Vo. At the same time,the output V3 of the first comparator 293B has H level since the voltageapplied to the positive input terminal of the first comparator 293B hasbeen raised to the second voltage Vr2. As a result, the first currentsource circuit 281 of the current source circuit 28B is activated.Hence, a voltage is generated across the offset resistor device 27 bythe current of the first current source circuit 281. The detectionvoltage Vs generated across the current detection resistor 26 has avalue lower than the voltage proportionate to the current flowing in theswitching device 21 by a predetermined value. In other words, althoughthe upper limit setting value Vc remains unchanged from the originalvalue, since the detection voltage Vs becomes lower, the ON time of theswitching device 21 becomes longer so that the detection voltage Vsfollows the upper limit setting value Vc. Eventually, the inductancecurrent IL becomes larger than the maximum value obtained during theordinary operation, and the charging of the output capacitor 31 proceedsquickly. As the output voltage Vo becomes close to the target value setby the second voltage Vr2 of the reference voltage source 290, theoutput V3 of the first comparator 293B becomes L level. Consequently,the first current source circuit 281 is deactivated and does not passany current. For this reason, no offset voltage is generated across theoffset resistor device 27, and the detection voltage Vs generated acrossthe current detection resistor 26 returns to the voltage proportionateto the current flowing in the switching device 21. In other words, thedetection voltage Vs rises. As a result, the inductance current IL islimited to the maximum value obtained during the ordinary operation, andthe charging of the output capacitor 31 is in a state of being braked.Eventually, the upper limit setting value Vc lowers, and the outputvoltage Vo is stabilized to the target value set by the second voltageVr2 of the reference voltage source 290.

Next, the operation in the case that, for the purpose of abruptlylowering the output voltage Vo to still another target value, thereference voltage Vr is set so as to be lowered abruptly from the secondvoltage Vr2 to the first voltage Vr1 will be described below.

Since the reference voltage Vr has been changed from the second voltageVr2 to the first voltage Vr1, the error amplifier 292 lowers the upperlimit setting value Vc to lower the output voltage Vo. At the same time,the output V4 of the second comparator 294B has H level since thevoltage applied to the negative input terminal of the second comparator294B has been lowered to the first voltage Vr1. As a result, the secondcurrent source circuit 282 of the current source circuit 28B isactivated. Hence, a voltage is generated across the offset resistordevice 27 by the current of the second current source circuit 282. Thedetection voltage Vs generated across the current detection resistor 26has a value higher than the voltage proportionate to the current flowingin the switching device 21 by a predetermined value. In other words,although the upper limit setting value Vc remains unchanged from theoriginal value, since the detection voltage Vs becomes higher, the ONtime of the switching device 21 becomes shorter so that the detectionvoltage Vs follows the upper limit setting value Vc. Eventually, theinductance current IL becomes smaller than the minimum value obtainedduring the ordinary operation and flows reversely. As a result, thedischarging of the output capacitor 31 proceeds quickly. As the outputvoltage Vo becomes close to the target value set by the first voltageVr1 of the reference voltage source 290, the output V4 of the secondcomparator 294B becomes L level. Consequently, the second current sourcecircuit 282 is deactivated and does not pass any current. For thisreason, no offset voltage is generated across the offset resistor device27, the detection voltage Vs generated across the current detectionresistor 26 returns to the voltage proportionate to the current flowingin the switching device 21, and the detection voltage Vs lowers. As aresult, the inductance current IL is limited to the minimum valueobtained during the ordinary operation, and the discharging of theoutput capacitor 31 is in a state of being braked. Eventually, the upperlimit setting value Vc rises, and the output voltage Vo is stabilized tothe target value set by the first voltage Vr1 of the reference voltagesource 290.

FIG. 10 is a characteristic graph showing the relationship between theinductance current IL and the detection voltage Vs in the switchingpower supply in accordance with the sixth embodiment. During theordinary operation, both the output V3 of the first comparator 293B andthe output V4 of the second comparator 294B have L level, and both thefirst current source circuit 281 and the second current source circuit282 of the current source circuit 28B are inactive. For this reason, nooffset voltage is generated, and the detection voltage Vs generatedacross the current detection resistor 26 is proportionate to the currentflowing in the switching device 21. When the output V3 of the firstcomparator 293B or the output V4 of the second comparator 294B becomes Hlevel to change the output voltage Vo, the offset voltage of the currentdetection circuit in the switching power supply is generated.

As shown in FIG. 10, in the switching power supply in accordance withthe sixth embodiment, the relationship between the inductance current ILand he detection voltage Vs is shifted back and forth from the zerovalue of the inductance current IL, whereby power supply andregeneration can be carrying out quickly. As a result, the switchingpower supply in accordance with the sixth embodiment is capable ofquickly coping with abrupt change in the output voltage and also capableof carrying out smooth return to the ordinary operation.

Although a step-down converter serving as a switching power supply hasbeen described in the fourth embodiment to the sixth embodimentdescribed above, the current detection circuit in accordance with thepresent invention is applicable not only to the step-down converter butalso to almost all switching power supplies. For example, a currentdetection circuit including a switching device connected to the powersupply potential side as in the first embodiment and the secondembodiment can detect the currents of high-side switches included instep-down converters, inverse converters and bridge converters.Furthermore, a current detection circuit including a switching deviceconnected to the ground potential side as in the third embodiment candetect the currents of low-side switches included in step-up converters,single-switch isolated converters and bridge converters.

As described above, the current detection circuit in accordance with thepresent invention, regardless of whether the circuit is applied to anytype of converters, produces excellent effects in the switching powersupplies described in the fourth embodiment to the sixth embodiments

Although an example wherein a switching device is formed of a MIS FET isdescribed in each embodiment described above, if a J-FET is used as aswitching device, a similar effect is produced.

The current detection circuit in accordance with the present inventioncan detect the current flowing in the negative direction in atransistor, such as a MIS FET. Hence, a switching power supply usingthis current detection circuit is useful as a power supply capable ofstabilizing the output voltage thereof in quick response to abruptchange in output conditions.

Although the invention has been described in its preferred form with acertain degree of particularity, it is understood that the presentdisclosure of the preferred form has been changed in the details ofconstruction and the combination and arrangement of parts may beresorted to without departing from the spirit and the scope of theinvention as claimed.

1. A current detection circuit for detecting the current flowing in aswitching device, comprising: an auxiliary switching device beingprovided in parallel with said switching device and being ON at leastwhen said switching device is ON, an offset voltage source forgenerating an offset voltage, and a compensation circuit for adjustingthe output current of said auxiliary switching device so that thepotential obtained by adding said offset voltage to the output potentialof said switching device becomes equal to the output potential of saidauxiliary switching device, wherein said offset voltage source comprisesan offset resistor device, and a current source circuit for passing acurrent in said offset resistor device, and said offset resistor deviceis an FET having the same channel type as that of said switching device.2. The current detection circuit in accordance with claim 1, whereinsaid switching device and said auxiliary switching device are FETshaving the same channel type.
 3. The current detection circuit inaccordance with claim 1, wherein said compensation circuit comprises: adifferential amplifier to which the potential obtained by adding saidoffset voltage to the output potential of said switching device and theoutput potential of said auxiliary switching device are input, and avariable impedance device, connected to the output terminal of saidauxiliary switching device, the impedance of which changes depending onthe output of said differential amplifier.
 4. A switching power supplycomprising a switching device, an inductor connected to said switchingdevice, a rectifier circuit for rectifying the output of said inductor,smoothing means for smoothing the current flowing in said rectifiercircuit, a current detection circuit for detecting the current of saidswitching device, and a control circuit, to which the current detectionsignal from said current detection circuit and the output voltage outputfrom said smoothing means are input, for turning ON/OFF said switchingdevice to control said output voltage, wherein said current detectioncircuit comprises: an auxiliary switching device being provided inparallel with said switching device and being ON at least when saidswitching device is ON, an offset voltage source for generating anoffset voltage, and a compensation circuit for adjusting the outputcurrent of said auxiliary switching device so that the potentialobtained by adding said offset voltage to the output potential of saidswitching device becomes equal to the output potential of said auxiliaryswitching device, wherein said offset voltage source comprises an offsetresistor device, and a current source circuit for passing a current insaid offset resistor device, and said offset resistor device is an FEThaving the same channel type as that of said switching device.
 5. Theswitching power supply in accordance with claim 4, wherein said controlcircuit generates a control signal for controlling the output voltageand turns ON/OFF said switching device so that the current detectionsignal from said current detection circuit follows said control signal.6. The switching power supply in accordance with claim 5, wherein saidrectifier circuit has a rectifier switching device, and said controlcircuit turns ON/OFF said switching device and said rectifier switchingdevice alternately.